1. Technical Field
This invention is in the field of radiant energy sensitive photocell circuits. More particularly, the invention provides a means for automatically compensating for quiescent current produced by a photodetector while amplifying the signal current.
2. Background and Description of the Prior Art
Photodetectors are made using a variety of mechanisms and materials. Chief classes of these are photoemissive devices, such as phototubes and photomultipliers, photoconductive resistors made from, e.g., cadmium sulfide, amorphous silicon, lead salts, etc., photovoltaic devices made from, e.g., silicon, indium arsenide, etc. and phototransistors, usually made from silicon. Photovoltaic devices are unique in that they will produce a signal current without the aid of an external power source (e.g., silicon solar cells). All others require an external bias voltage. As a result, even in the absence of radiation, a so-called dark current will flow. Since some background radiation is always present, the term quiescent current is more accurate but the terms are often used interchangeably. Sometimes it is desirable to measure the background radiation. Herein, the term quiescent current means that caused by sources which are not desired to be measured directly as opposed to signal currents.
The quiescent current for photoemissive and photovoltaic devices is, in general, smaller than for photoconductive devices and also, as a general rule, devices using materials which are sensitive to longer wavelength radiation have higher quiescent currents. In fact, for photoconductors used to detect infrared radiation, the quiescent current is orders of magnitude larger than the usually obtained signal current. Thus, circuits which are suitable for other classes of photocells are usually inadequate for photoconductors. Nonetheless, assuming suitable circuitry, infrared photoconductors such as PbS and PbSe possess several advantages in the detection of infrared radiation. Among these are high sensitivity, room temperature operation, low cost, and ease of fabrication as high density arrays.
Photoconductors produce a signal by a variation in conductance roughly proportional to the intensity of infrared radiation. As mentioned, this conductance change is very small compared to the conductance when no radiation is incident. When a fixed external voltage is applied across the photoconductor the resulting current may be described as having two components, the quiescent current due to the quiescent conductance and the much smaller photo current due to the infrared radiation induced conductance change. Amplifying a small signal current in the presence of a large fixed current is difficult and with photoconductors it is even more so because the quiescent conductance varies with the temperature of the detector and there may be a significant component due to variations in any background radiation. Thus, photoconductors require carefully conceived signal processing techniques to maintain their advantages in a commercially viable system.
For photoconductors, if the optical signal is constant or varies at a slow rate, the most favored method for separating the photo current from the quiescent current is to modulate the optical signal, usually with a mechanical chopper. This modulation translates the information in the optical signal to the chopping frequency. The photo current may then be separated from the quiescent current by some form of high pass filtering, wherein signals at frequencies below the chopping frequency are attenuated while signals at or above the chopping frequency are amplified.
In systems using a small number of detectors, high pass filtering is usually accomplished using discrete capacitors and resistors. A simple well-known basic circuit is illustrated in FIG. 1. The photoconductor is biased with voltages, E+ and E-, through a bias resistor, Rb. The bias voltages may be of any polarity as long as there is a difference between them. When the resistance of the photodetector changes, the voltage at the junction with Rb changes and is transmitted by the coupling capacitor Cc to the amplifier A with input resistance Rin. If Rin is large compared to Rd or Rb, the high pass filter cut on frequency is determined by the inverse of the product of Cc and Rin.
For lead salt detectors, the desirable gain of the amplifier is typically on the order of as much as one million. However, with one or two operational amplifiers, it is easy to achieve such high gains as well as large input resistances. Since lead salt detectors work best with chopping frequencies below about one kilohertz, a high input resistance reduces the need for large value coupling capacitors. Even so, if they were required, in discrete circuits, capacitor size is not a major problem.
However, there are a number of applications such as spectroscopic instruments and imaging devices which require a large numbers of detectors. An emerging standard for low cost infrared spectroscopic instruments is a 128 detector linear array. For scanned imaging systems, 512 or more is preferable and there are applications where two dimensional arrays with thousands of detectors are desired. In these applications, integrated circuits with a large number of detector amplifier channels per chip are preferred. However, operational amplifiers and large value capacitors would take up too much chip area. A typical instrumentation op amp might be 0.01 sq. in. and 32 channels of these would require 0.32 sq. in. for these alone. This is too large to be economical enough for most applications. Thus, different approaches are required.
Referring back to FIG. 1, one possibility is to make the bias voltages equal and opposite and the bias resistor the same value as the quiescent resistance of the photodetector. In this case, the DC voltage at the junction is zero and the coupling capacitor can be eliminated. However, this is not a workable solution. As noted, the amplifier gain is so large that, in practice, changes in photodetector resistance due to temperature changes or background radiation quickly unbalance the circuit so the output of the amplifier saturates. For lead salt detectors, the temperature coefficient of resisitance is about -2%/.degree.C. so that small temperature changes are troublesome. Using a bridge circuit with a matched photodetector and a differential amplifier is a well-known improvement. Unfortunately, this reduces the signal to noise and roughly doubles the number of components.
One step known in the prior art which has been taken toward solving the problem of providing amplifier gain using small amounts of chip area is to integrate the current from the photodetector as illustrated in FIG. 2. In this circuit, the MOSFET transistor Q serves as an impedance transformer where the resistance presented to the detector is low (hundreds of k.OMEGA.) relative to the resistance presented to the capacitor Ci (thousands of M.OMEGA.). Also, if the amplifier a has a MOSFET input, Rin will be large enough to be ignored. Thus, Ci is fed by a current source with value I.sub.net =I.sub.det -I.sub.b. In operation, the switch, S, is closed until the integrating capacitor is discharged (reset) and then opened under the control of RST. The voltage on Ci is simply V.sub.i =(1/Ci).intg.I.sub.net (t) dt. For the purposes of illustration, assume that I.sub.net is constant so that V.sub.i =I.sub.net .times.T.sub.i /Ci. With typical values of integration time T.sub.i =1 ms and Ci=10 pf (which takes up little chip area), T.sub.i /Ci=100 M.OMEGA.. In the circuit of FIG. 1, with typical values of Rdet=Rb=10 M.OMEGA., the parallel resistance R.sub.p =5 M.OMEGA. so that a relative voltage gain of 20 is obtained by the circuit of FIG. 2. Higher gains can be obtained by reducing Ci or increasing T.sub.i.
An initial gain of 20 or more is enough to overcome the effects of any noise that might be introduced by later signal processing. In particular, the amplifier following Ci can be a simple buffer and it becomes possible to delay further amplification until the multiple signals have been sampled and held, passed through a multiplexor, and all channels amplified in sequence by a single additional amplifier residing off the chip.
Unfortunately, the drawback to using a reset current integrator is that it amplifies the, often much larger, DC quiescent current as well and can result in saturation of the following amplifier. Besides chip area, selecting a value for Ci always results in a tradeoff between having enough signal gain and avoiding saturation. For each application, the signal frequency will determine T.sub.i and the detector characteristics will determine the net DC current I.sub.net into Ci. For detectors with large quiescent currents, to avoid saturation, the bias current I.sub.b must still be matched to the quiescent current I.sub.det with a precision which is roughly better than the inverse of the gain (5% in the example). As noted above, this is hard to accomplish and/or maintain as ambient conditions change, especially for a large number of channels. An attempt to solve this problem for lead salt detectors, hopefully allowing for use without any intentional chopping, which is commercially available is believed to be illustrated in FIG. 3.
This differs from the previous prior art example in that, instead of a fixed bias resistor acting as a balancing current sink, MOSFET transistor M3 in conjunction with an on-chip digital-to-analog convertor D/A act as a variable current sink. Transistors M1 and M2 are voltage level translators, but have no intentional effect on the currents charging the integration capacitor CW. The amount of current required from the D/A is determined by a calibration step in which the detectors are exposed to a DC radiation source, the detector current is integrated by CW, sampled and held by S/H, and converted to a digital value by the off-chip analog-to-digital convertor A/D CONVERTOR. Off-chip DIGITAL PROCESSING then calculates a digital value which is sent to the on-chip digital STORAGE REGISTER which drives the D/A.
This approach works under some conditions but has several disadvantages. The major one is that the temperature coefficient of resistance of the photodetectors is so large that the quiescent current will drift large amounts even with reasonable temperature control. Thus, signal measurements must be interrupted and the calibration step preformed from time to time. Even so, temperature controllers are expensive, bulky, and consume many times the power of an integrated circuit. Second, every photodetector must have its own associated on-chip digital-to-analog convertor. This requires a large amount of chip area unless the resolution of the convertor is very low. If the resolution is low, then it will be difficult to match the photodetector quiescent current and, therefore, the gain at the integration capacitor must be reduced to avoid saturation. Lastly, in some situations it may not be convenient or cost effective to require an external analog-to-digital converter and digital processor.